Abstract
This application note provides guidelines and suggestions for RF printed-circuit board (PCB) design and layout, including some discussion of mixed-signal applications. The material provides "best practices" guidance, and should be used in conjunction with all other design and manufacturing guidelines that may apply to particular components, PCB manufacturers, and material sets as applicable.
This application note applies to all Analog Device wireless products.
Introduction
This application note provides guidelines and suggestions for RF printed-circuit board (PCB) design and layout, including some discussion of mixed-signal applications, such as digital, analog, and RF components on the same PCB. The material is arranged by topic areas and provides "best practices" guidance. It should be used in conjunction with all other design and manufacturing guidelines that may apply to particular components, PCB manufacturers, and material sets as applicable.
RF Transmission Lines
Many of Analog Devices' RF components require controlled impedance transmission lines that will transport RF power to (or from) IC pins on the PCB. These transmission lines can be implemented on a exterior layer (top or bottom), or buried in an internal layer. Guidelines for these transmission lines include discussions relating to the microstrip, suspended stripline, coplanar waveguide (grounded), and characteristic impedance. It also describes transmission line bends and corner compensation, and layer changes for transmission lines.
Microstrip
This type of transmission line consists of fixed-width metal routing (the conductor), along with a solid unbroken ground plane located directly underneath (on the adjacent layer). For example, a microstrip on Layer 1 (top metal) requires a solid ground plane on Layer 2 (Figure 1). The width of the routing, the thickness of the dielectric layer, and the type of dielectric determine the characteristic impedance (typically 50Ω or 75Ω).
Figure 1. Microstrip example (isometric view).
Suspended Stripline
This line consists of a fixed-width routing on an inner layer, with solid ground planes above and below the center conductor. The conductor can be located midway between the ground planes (Figure 2), or it can be offset (Figure 3). This is the appropriate method for RF routing on inner layers.
Figure 2. Suspended stripline (end view).
Figure 3. Offset suspended stripline. A variant of the stripline, for PCBs with unequal layer thicknesses (end view).
Coplanar Waveguide (Grounded)
A coplanar waveguide provides for better isolation between nearby RF lines, as well as other signal lines (end view). This medium consists of a center conductor with ground planes on either side and below (Figure 4).
Figure 4. A coplanar waveguide provides for better isolation between nearby RF lines and other signal lines.
Via "fences" are recommended on both sides of a coplanar waveguide, as shown in Figure 5. This top view provides an example of a row of ground vias on each top metal gound plane on either side of the center conductor. Return currents induced on the top layer are shorted to the underlying ground layer.
Figure 5. Via fences are recommended on both sides of a coplanar waveguide.
Characteristic Impedance
There are several calculators available to properly set the signal conductor line width to achieve the target impedance. However, caution should be used when entering the dielectric constant of the layers. The outer laminated layers of typical PCBs often contain less glass content than the core of the board, and consequently the dielectric constant is lower. For example, FR4 core is generally given as εR = 4.2, whereas the outer laminate (prepreg) layers are typically εR = 3.8. Examples given below for reference only, metal thickness assumed for 1oz copper (1.4 mils, 0.036mm).
| Media | Dielectric | Layer Thickness in mils (mm) | Center Conductor in mils (mm) | Gap | Characteristic Impedance |
| Microstrip | Prepreg (3.8) | 6 (0.152) | 11.5 (0.292) | N/A | 50.3 |
| 10 (0.254) | 20 (0.508) | 50.0 | |||
| Diff. Pair | Prepreg (3.8) | 6 (0.152) | 25 (0.635) | 6 (0.152) | 50.6 |
| Stripline | FR4 (4.5) | 12 (0.305) | 3.7 (0.094) | N/A | 50.0 |
| Offset Stripline | Prepreg (3.9) |
6 (0.152) upper, | 4.8 (0.122) |
N/A |
50.1 |
| 10 (0.254) lower | |||||
| Coplanar WG | Prepreg (3.8) |
6 (0.152) | 14 (0.35) | 20 (0.50) | 49.7 |
Transmission Line Bends and Corner Compensation
When transmission lines are required to bend (change direction) due to routing constraints, use a bend radius that is at least 3 times the center conductor width. In other words:
Bend Radius ≥ 3 × (Line Width).
This will minimize any characteristics impedance changes moving through the bend.
In cases where a gradually curved bend is not possible, the transmission line can undergo a right-angle bend (noncurved). See Figure 6. However, this must be compensated to reduce the impedance discontinuity caused by the local increase in effective line width going through the bend. A standard compensation method is the angled miter, as illustrated below. The optimum microstrip right-angle miter is given by the formula of Douville and James:
Where M is the fraction (%) of the miter compared to the unmitered bend. This formula is independent of the dielectric constant, and is subject to the constraint that w/h ≥ 0.25.
Similar methods can be employed for other transmission lines. If there is any uncertainty as to the correct compensation, the bend should be modeled using an electromagnetic simulator if the design requires high-performance transmission lines.
Figure 6. When a curved bend is not possible, the transmission line can undergo a right-angle bend.
Layer Changes for Transmission Lines
When layout constraints required that a transmission line move to a different layer, it is recommended that at least two via holes be used for each transition to minimize the via inductance loading. A pair of vias will effectively cut the transition inductance by 50%, and the largest diameter via should be utilized that is compatible with the transmission line width. For example, on a 15-mil microstrip line, a via diameter (finished plated diameter) of 15 mils to 18 mils would be used. If space does not permit the use of larger vias, then three transition vias of smaller diameter should be used.
Signal Line Isolation
Care must be taken to prevent unintended coupling between signal lines. Some examples of potential coupling and preventative measures:
- RF Transmission Lines: Lines should be kept as far apart as possible, and should not be routed in close proximity for extended distances. Coupling between parallel microstrip lines will increase with decreasing separation and increasing parallel routing distance. Lines that cross on separate layers should have a ground plane keeping them apart. Signal lines that will carry high power levels should be kept away from all other lines whenever possible. The grounded coplanar waveguide provides for excellent isolation between lines. It is impractical to achieve isolation better than approximately -45dB between RF lines on small PCBs.
- High-Speed Digital Signal Lines: These lines should be routed separately on a different layer than the RF signal lines, to prevent coupling. Digital noise (from clocks, PLLs, etc.) can couple onto RF signal lines, and these can be modulated onto RF carriers. Alternatively, in some cases digital noise can be up/down-converted.
- VCC/Power Lines: These should be routed on a dedicated layer. Adequate decoupling/bypass capacitors should be provided at the main VCC distribution node, as well as at VCC branches. The choice of the bypass capacitances must be made based on the overall frequency response of the RF IC, and the expected frequency distribution nature of any digital noise from clocks and PLLs. These lines should also be separated from any RF lines that will transmit large amounts of RF power.
Ground Planes
The recommended practice is to use a solid (continuous) ground plane on Layer 2, assuming Layer 1 is used for the RF components and transmission lines. For striplines and offset striplines, ground planes above and below the center conductor are required. These planes must not be shared or assigned to signal or power nets, but must be uniquely allocated to ground. Partial ground planes on a layer, sometimes required by design constraints, must underlie all RF components and transmission lines. Ground planes must not be broken under transmission line routing.
Ground vias between layers should be added liberally throughout the RF portion of the PCB. This helps prevent accrual of parasitic ground inductance due to ground-current return paths. The vias also help to prevent cross-coupling from RF and other signal lines across the PCB.
Special Consideration on Bias and Ground Layers
The layers assigned to system bias (DC supply) and ground must be considered in terms of the return current for the components. The general guidance is to not have signals routed on layers between the bias layer and the ground layer.
Figure 7. Incorrect layer assignment: there are signal layers between the bias layer and ground-current return path on ground layer. Bias line noise can be coupled to the signal layers.
Figure 8. Better layer assignment: there are no signal layers between the bias and ground return layers.
Power (Bias) Routing and Supply Decoupling
A common practice is to use a "star" configuration for the power-supply routes, if a component has several supply connections (Figure 9). A larger decoupling capacitor (tens of µFds) is mounted at the "root" of the star, and smaller capacitors at each of the star branches. The value of these latter capacitors depends on the operating frequency range of the RF IC, and their specific functionality (i.e., interstage vs. main supply decoupling). An example is shown below.
Figure 9. If a component has several supply connections, the power-supply routes can be arranged in a star configuration.
The "star" configuration avoids long ground return paths that would result if all the pins connected to the same bias net were connected in series. A long ground return path would cause a parasitic inductance that could lead to unintended feedback loops. The key consideration with supply decoupling is that the DC supply connections must be electrically defined as AC ground.
Selection of Decoupling or Bypass Capacitors
Real capacitors have limited effective frequency ranges due to their self-resonant frequency (SRF). The SRF is available from the manufacturer, but sometimes must be characterized by direct measurement. Above the SRF, the capacitor is inductive, and therefore will not perform the decoupling or bypass function. When broadband decoupling is required, standard practice is to use several capacitors of increasing size (capacitance), all connected in parallel. The smaller value capacitors normally have higher SRFs (for example, a 0.2pF value in a 0402 SMT package with an SRF = 14GHz), while the larger values have lower SRFs (for example, a 2pF value in the same package with an SRF = 4GHz). A typical arrangement is depicted in Table 2.
| Component | Capacitance | Package | SRF | Useful Frequency Range* |
| Ultra-High Range | 20pF | 0402 | 2.5GHz | 800MHz to 2.5GHz |
| Very High Range | 100pF | 0402 | 800MHz | 250MHz to 800MHz |
| High Range | 1000pF | 0402 | 250MHz | 50MHz to 250MHz |
| Midrange | 1µF | 0402 | 60MHz | 100kHz to 60MHz |
| Low Range | 10µF | 0603 | 600kHz | 10kHz to 600kHz |
| *Low end of useful frequency range defined as less than 5Ω of capacitive reactance. | ||||
Bypass Capacitor Layout Considerations
Since the supply lines must be AC ground, it is important to minimize the parasitic inductance added to the AC ground return path. These parasitic inductances can be caused by layout or component orientation choices, such as the orientation of a decoupling capacitor's ground. There are two basic methods, shown in Figure 10 and Figure 11.
Figure 10. This configuration presents the smallest total footprint for the bypass capacitor and related vias.
In this configuration, the vias connecting the VCC pad on the top layer to the inner power plane (layer) potentially impede the AC ground current return, forcing a longer return path with resulting higher parasitic inductance. Any AC current flowing into the VCC pin passes through the bypass capacitor to its ground side before returning on the inner ground layer. This configuration presents the smallest total footprint for the bypass capacitor and related vias.
Figure 11. This configuration requires more PCB area.
In this alternate configuration, the AC ground return paths are not blocked by the power-plane vias. Generally this configuration requires somewhat more PCB area.
Grounding of Shunt-Connected Components
For shunt-connected (grounded) components (such as power-supply decoupling capacitors), the recommended practice is to use at least two grounding vias for each component (Figure 12). This reduces the effect of via parasitic inductance. Via ground "islands" can be used for groups of shunt-connected components.
Figure 12. Using at least two grounding vias for each components reduces the effect of via parasitic inductance.
IC Ground Plane ("Paddle")
Most ICs require a solid ground plane on the component layer (top or bottom of PCB) directly underneath the component. This ground plane will carry DC and RF return currents through the PCB to the assigned ground plane. The secondary function of this component "ground paddle" is to provide a thermal heatsink, so the paddle should include the maximum number of thru vias that are allowed by the PCB design rules. The example below shows a 5 × 5 array of via holes embedded in the central ground plane (on the component layer) directly under the RF IC (Figure 13). The maximum number of vias that can be accommodated by other layout considerations should be used. These vias are ideally thru-vias (i.e., penetrate all the way through the PCB), and must be plated. If possible, the vias should be filled with thermally conductive paste to enhance the heatsink (the paste is applied after via plating and prior to final board plating).
Figure 13. A 5 × 5 array of via holes embedded in the central ground plane directly under the RF IC.
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High-Dynamic-Range, Direct Up-/Downconversion 1500MHz to 2500MHz Quadrature Mod/Demod
Low-Cost RF Up/Downconverter with LNA and PA Driver
Integrated L1-Band GPS Receiver
Scalable High-Density Downstream Cable QAM Modulator
1300MHz to 2700MHz Variable-Gain Amplifier with Analog Gain Control
40MHz to 4GHz Linear Broadband Amplifiers
AISG Integrated Transceiver
High-Gain Vector Multipliers
2MHz to 1.6GHz 45dB RF-Detecting Controllers and RF Detector
Complete, Direct-Conversion Tuner for MMDS Applications
Adjustable, High-Linearity, SiGe, Dual-Band, LNA/Mixer ICs
RF Power Detector
3.5GHz Downconverter Mixers with Selectable LO Doubler
5GHz, 5-Channel MIMO Receiver
+2.7V, Single-Supply, Cellular-Band Linear Power Amplifiers
900MHz Image-Reject Transceivers
VHF-to-Microwave, +3V, General-Purpose Amplifiers
Single-Conversion DVB-H Tuner
+3.6V, 1W Autoramping Power Amplifier for 900MHz Applications
3.6V, 1W RF Power Transistors for 900MHz Applications
Low-Power, Cellular Upconverter-Driver
W-CDMA LNA/Mixer ICs
Tiny Low-Noise Amplifiers for HSPA/LTE
3.3GHz to 3.9GHz Wireless Broadband RF Transceiver
Upstream CATV Amplifier
1575MHz/1900MHz Variable-IP3 Low-Noise Amplifiers
+14dBm to +20dBm LO Buffers/Splitters with ±1dB Variation
315MHz/433MHz ASK Superheterodyne Receiver with Extended Dynamic Range
500MHz to 4000MHz Dual Analog Voltage Variable Attenuator with On-Chip 10-Bit SPI-Controlled DAC
W-CDMA/W-TDD/TD-SCDMA Zero-IF Receivers
1.2GHz VCO with Linear Modulation Input
2.4GHz to 2.5GHz 802.11g/b RF Transceiver with PA and Rx/Tx/Diversity Switch
Complete Dual-Band Quadrature Transmitters
+14dBm to +20dBm LO Buffers with ±1dB Variation
50MHz to 1000MHz Analog VGA with Threshold Alarm Circuit and Error Amplifier for Level Control
Low-Cost, Crystal-Based, Programmable, ASK/FSK Transceiver with Fractional-N PLL
900MHz SiGe, High-Variable IP3, Low-Noise Amplifier
600mA/650mA PWM Step-Down Converters in 2mm x 2mm WLP for WCDMA PA Power
SiGe, High-Linearity, High-Gain, 2000MHz to 3000MHz Downconversion Mixer with LO Buffer
High-Dynamic-Range, Direct Up/Downconversion 1500MHz to 3000MHz Quadrature Modulator/Demodulator
Complete, Direct-Conversion Tuner for DVB-S and Free-to-Air Applications
1.7GHz to 2.5GHz, Direct I/Q Modulator with VGA and PA Driver
Low-Voltage IF Transceiver with Limiter and RSSI
40MHz to 4GHz Linear Broadband Amplifiers
High-Gain Vector Multipliers
45MHz to 650MHz, Integrated IF VCOs with Differential Output
2.4GHz Monolithic Voltage-Controlled Oscillator
23.5MHz to 6000MHz Fractional/Integer-N Synthesizer/VCO
Complete DBS Direct-Conversion Tuner ICs with Monolithic VCOs
Triple/Dual-Mode CDMA LNA/Mixers
Complete Single-Conversion Television Tuner
DBS Direct Downconverter
RF Power Detectors in UCSP
400MHz to 2.5GHz, Low-Noise, SiGe Downconverter Mixers
500MHz to 2500MHz, VCO Buffer Amplifiers
2.7V, Single-Supply, Cellular-Band Linear Power Amplifiers
Global Automotive TV Tuner
FM Automotive Low-Noise Amplifier
900MHz Image-Reject Transceivers
VHF-to-Microwave, +3V, General-Purpose Amplifiers
Direct-Conversion to Low-IF Tuners for Digital Audio Broadcast
Direct-Conversion TV Tuner
Dual 50MHz to 1000MHz High-Linearity, Serial/Parallel-Controlled Digital VGA
200mW Single-Chip Transmitter ICs for 868MHz/915MHz ISM Bands
Dual, SiGe High-Linearity, High-Gain, 1800MHz to 2900MHz Downconversion Mixer with LO Buffer/Switch
Low-Noise, High-Linearity Broadband Amplifier
SiGe, High-Linearity, 2000MHz to 3900MHz Downconversion Mixer with LO Buffer
Upstream CATV Amplifiers
GPS/GNSS Ultra-Low-Noise-Figure LNAs
2.3GHz to 2.7GHz Wireless Broadband RF Transceiver
Band II and V WCDMA Femtocell Transceiver with GSM Monitoring
Complete Dual-Band Quadrature Transmitters
1575MHz/1900MHz Variable-IP3 Low-Noise Amplifiers
2.4GHz/5GHz, Single-Band and Dual-Band, 802.11g/a RF Transceiver ICs
SiGe High-Linearity, 815MHz to 995MHz Downconversion Mixer with LO Buffer/Switch
3.4GHz to 3.8GHz SiGe Low-Noise Amplifier/PA Predriver
2.4GHz to 2.5GHz Linear Power Amplifier
FM Automotive Low-Noise Amplifier
ISDB-T 1- and 3-Segment Low-IF Tuners
Dual, SiGe High-Linearity, 3000MHz to 4000MHz Downconversion Mixer with LO Buffer
CDMA IF VGAs and I/Q Demodulators with VCO and Synthesizer
300MHz-to-450MHz Low-Power, Crystal-Based ASK Transmitter
ISDB-T 1- and 3-Segment Low-IF Tuners
Femto Basestation Bits-to-RF Radio Transmitter
Low-Cost Downconverter with Low-Noise Amplifier
W-CDMA/W-TDD/TD-SCDMA Zero-IF Receivers
400MHz to 2.5GHz Upconverters
1200MHz to 2500MHz Adjustable RF Predistorter
Advanced Multimode Complete RF-to-Baseband Receiver
400MHz to 2.5GHz Upconverters
ISDB-T Single-Segment Low-IF Tuners
900MHz Image-Reject Transceivers
Adjustable, High-Linearity, SiGe, Dual-Band, LNA/Mixer ICs
Dual, SiGe, High-Linearity, High-Gain, 700MHz to 1000MHz Downconversion Mixer with LO Buffer/Switch
WLAN/WiMAX Low-Noise Amplifiers
Low-Cost, 900MHz, Low-Noise Amplifier and Downconverter Mixer
315MHz/390MHz Dual-Frequency ASK Transmitter
Complete RF-to-Baseband Receiver
315MHz/433MHz Low-Noise Amplifier for Automotive RKE
50MHz to 1000MHz High-Linearity, Serial/Parallel-Controlled Analog/Digital VGA
High-Density Downstream Cable QAM Modulator
1700MHz to 2200MHz, High-Linearity, SPI-Controlled DVGA with Integrated Loopback Mixer
Low-Voltage IF Transceiver with Limiter RSSI and Quadrature Modulator
Complete Cellular Baseband-to-RF Transmitters with PA
Complete Single-Conversion Television Tuner
Direct-Conversion Tuner ICs for Digital DBS Applications
+2.7V, Single-Supply, Cellular-Band Linear Power Amplifiers
+14dBm to +20dBm LO Buffers with ±1dB Variation
Quad-Band TDD-WCDMA RF-to-Bits Radio Receiver
Satellite IF Switch
Direct-Conversion to Low-IF Tuners for Digital Audio Broadcast
10MHz to 1050MHz Integrated RF Oscillator with Buffered Outputs
2.5GHz, 22dBm/20dBm Power Amplifiers with Analog Closed-Loop Power Control
GPS/GNSS Low-Noise Amplifiers with Integrated LDO
Dual 50MHz to 1000MHz High-Linearity, Serial/Parallel-Controlled Analog/Digital VGA
WCDMA Quasi-Direct Modulator with VGA and PA Driver
CDMA IF VGAs and I/Q Demodulators with VCO and Synthesizer
W-CDMA LNA/Mixer ICs
GPS/GNSS LNAs with Antenna Switch and Bias
RMS Power Detector
Tiny Low-Noise Amplifiers for HSPA/LTE
2.4GHz to 2.5GHz, 802.11g RF Transceivers with Integrated PA
Upstream CATV Amplifier
Band I, V, and VIII WCDMA Femtocell Transceiver with GSM Monitoring
Complete Dual-Band Quadrature Transmitters
GPS/GNSS Low-Noise Amplifiers
2.4GHz/5GHz, Single-Band and Dual-Band, 802.11g/a RF Transceiver ICs
1575MHz GPS Front-End Amplifier
2.4GHz SiGe, High IP3 Low-Noise Amplifier
Dual RF LDMOS Bias Controllers with I²C/SPI Interface
2.4GHz 802.11b Zero-IF Transceivers
SiGe, High-Linearity, 2300MHz to 4000MHz Downconversion Mixer with LO Buffer
50MHz to 1000MHz, 75dB Logarithmic Detector/Controller
Quad-Band TDD-WCDMA Bits-to-RF Radio Transmitter
10MHz to 500MHz Dual Analog Voltage Variable Attenuator with On-Chip 10-Bit SPI-Controlled DAC
Complete Cellular Baseband-to-RF Transmitter with PA
Dual, SiGe, High-Linearity, High-Gain, 700MHz to 1000MHz Downconversion Mixer with LO Buffer/Switch
W-CDMA/W-TDD/TD-SCDMA Zero-IF Receivers
400MHz to 2.5GHz Upconverters
DC-to-Microwave, +5V Low-Noise Amplifier
700MHz to 2700MHz Analog VGA with Threshold Alarm Circuit and Error Amplifier for Level Control
315MHz/434MHz ASK Superheterodyne Receiver
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