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Download this article in PDF format. (669KB) High-Side Current Sensing with Wide Dynamic Range: Three Solutions
To understand the key trade-offs, options, and challenges faced by system designers when choosing the most accurate, cost-effective current sensor for a circuit board, we take a close look at current sensing in LDMOS bias current monitoring in cellular base station power amplifiers and other relevant applications. Current
monitoring is necessary in base station power amplifiers, especially
with the more complex modulation methods used in 3G and LTE,
where the peak-to-average power ratio varies from 3.5 dB (about
2.2 to 1) for 3G W-CDMA to 8.5 dB (about 7.1 to 1) for LTE OFDM—compared
with 3 dB (about 2 to 1) for the most popular 2G single-carrier
GSM. One of the control-loop functions is to monitor the LDMOS
bias current, which allows the bias of the LDMOS to be properly
modulated for a given power output. Typically, this dc bias
current has a wide dynamic range based on operating, maximum,
or off-peak operation. To the designer, this means that an accurate
current sensor is needed to monitor a current that can range
from 50 mA (or as little as 15 mA) While shunt resistors are available to handle this power, lower power dissipation might be a requirement of the board. But the choice of such low resistance values means that at low currents, say 50 mA, the voltage across the 10 mΩ shunt is extremely small (500 μV), making it a challenge to monitor accurately—with a circuit that must also withstand high common-mode voltage. This article will focus on providing current-sensing solutions that can help designers accurately monitor wide-ranging dc currents in the presence of high common-mode voltages. Special attention will also be devoted to temperature performance, a critical parameter that is often not easy to calibrate but must be faced in the case of outdoor power amplifiers. Three optional solution approaches are described here—in order of decreasing design complexity—that provide viable high-accuracy, high-resolution current sensing for a variety of applications. - Use discrete components, such as op amps, resistors, and Zener diodes, to build a current sensor. This solution employs the AD8628 zero-drift amplifier as its key component.
- Attain an increased level of integration by using a high voltage bidirectional current shunt monitor, such as the AD8210, with additional external components to extend dynamic range and accuracy.
- Employ an application-optimized device such as the newly available AD8217, an easy-to-use, highly integrated zero-drift current sensor with an input common-mode voltage range of 4.5 V to 80 V.
This
circuit monitors the high-side current, In
this configuration, the voltage across R because
feedback via the MOSFET maintains both high-impedance op-amp
inputs at the same voltage. The current through _{SHUNT}R
flows through the FET and _{G}R to develop _{L}V.
The relationship between the current, _{OUTPUT}I, flowing through
the shunt resistor and V is expressed
by Equation 1:_{OUTPUT}
R
is limited by the allowable power consumption at maximum current.
The minimum value of _{SHUNT}R is limited by
the input range and error budget of the op amp. Normally, the
value of _{SHUNT}R is from 1 mΩ
to 10 mΩ
for monitoring currents greater than 10 A. If a single resistor
cannot meet the power consumption requirement or is too large
for the PCB, _{SHUNT}R may have to be made up
of multiple resistors in parallel. _{SHUNT}
R is used to translate a current
proportional to the high-side current to the low-side. The maximum
_{G}R is limited by the drain-source leakage
current of the P-channel MOSFET. For example, consider the common
P-channel enhancement-mode vertical DMOS transistor BSS84. The
maximum _{G}I at various conditions is shown
in Table 1._{DSS}
Consider
the LDMOS drain current monitoring example, with 28-V common
mode and R should be at least
20 times _{L}I. This results in_{DSS}
The
minimum
R divides
to produce the op amp's quiescent current and the essentially
constant Zener-diode voltage, _{BIAS}V, (which determines
the op amp's supply voltage). Make sure the current flowing
through the Zener diode does not exceed its maximum regulated
current, _{Z}I, when the amplifier current,
_{Z_MAX}I, is essentially zero and _{SUPPLY}V
is the maximum:_{IN}
To
ensure a stable diode voltage, the current flowing through it
should be higher than its minimum operating current, I is the maximum and _{SUPPLY}V
is the minimum:_{IN}
The
Zener diode and
R_{1} is used to limit the amplifier
input current if input transients exceed the op amp's power
supply voltage. A The
offset voltage, I,
of the chosen op amp are critical, especially with low values
of shunt resistance and at low load currents. _{OS}V
+ _{OS}I
× _{OS}R_{1} must be smaller than I
× _{MIN}R, or the amplifier may be saturated.
Therefore, a rail-to-rail input amplifier with zero crossover
distortion is preferred for optimal performance._{SHUNT}Another
issue to consider for this discrete solution is temperature
drift. Even if an amplifier with zero drift is used, it is very
difficult or costly to optimize drifts caused by discrete components:
the Zener diode, MOSFET, and resistors. From Table 1, the maximum
V = 0 V and _{GS}V =
–50 V. This drift will degrade the system accuracy over
temperature, especially when the monitored current is low. The
drift of the Zener diode will impact the stability of the amplifier's
power supply, so the amplifier used should have high power-supply
rejection (PSR). _{DS}Further,
designers must take into account the low power efficiency of
this solution: A lot of power is consumed by R is a 1000-Ω resistor, the circuit will dissipate
more than 0.52 W of undesirable power. This adds to the power-consumption
budget and must be taken into account._{BIAS}
The AD8210 amplifies a small differential input voltage generated by a positive or negative current flowing through the shunt resistor. The AD8210 rejects high common-mode voltages (up to 65 V) and provides a ground-referenced buffered output. As
shown in Figure 2a, it comprises two main blocks, a differential
amplifier, and an instrumentation amplifier. The input terminals
are connected to differential amplifier A1 via R The
differential currents through Q1 and Q2 are converted into a
differential voltage by R The
output reference voltage is easily adjusted using the V This
configuration works well for charge/discharge applications,
but if the user needs to utilize the entire output range to
measure a unidirectional current flow, then the circuit of Figure
2b shows a typical way that an external source may be used to
set the range. Here a resistive divider is buffered by an op
amp to drive the V It is difficult for the amplifier alone to monitor the load current as it comes close to zero. With a 5-V supply, the AD8210 specifies a linear output range with a minimum output of 50 mV and a maximum output of 4.9 V. Consider an application where the shunt resistance is 10 mΩ. The minimum current through it must be greater than 250 mA to ensure that the output of the AD8210 is above its lowest point of 50 mV.
The configuration shown in Figure 2b adds an offset to allow smaller currents to be measured. The relationship between output voltage and monitor current, based on an amplifier gain of 20 V/V, can be calculated as Equation 2:
For
example, using resistances, In fact, using this configuration, the designer can offset the output of the AD8210 to any point within its supply range in order to handle arbitrary current ranges having any degree of asymmetry. An op amp to buffer the voltage divider is desirable because precision-trimmed resistances are connected internally to the reference inputs—so, for best results, those inputs should be driven at low impedance. Precision and low cost op amps that can be used to buffer the external reference include—for example—the AD8541, AD8601, AD8603, AD8605, AD8613, AD8691, and AD8655. Compared
with the discrete solution, this integrated solution requires
that the current shunt monitor have high common-mode voltage
range and an output offset—if the output voltage range
cannot meet the current-detection range requirements. But it
can handle bidirectional current monitoring, and it avoids the
temperature drift and power consumption issues described above.
The offset drift and gain drift of the AD8210 are guaranteed
to be a maximum of 8 µV/°C and 20 ppm/°C, respectively.
And if, for example, the AD8603 were used as the buffer, it
would contribute an offset of only 1 µV/°C, which
can be neglected compared with the AD8210's already low offset-voltage
drift. The power consumption of the divider, R
or only 1.2 mW, using the parameters in Figure 2b.
For measuring very small currents through a small shunt resistor, the AD8217 features a minimum 20-mV output range over temperature, an improvement over the AD8210's 50-mV range. Thus, if the minimum load current being monitored across the shunt produces 20-mV minimum output, which is 1-mV minimum input, from the current sensor, then the user can choose the AD8217 configured as shown in Figure 3b. The relationship between the output voltage of the AD8217 and the input current can be calculated as Equation 3:
The
AD8217 features an internal The AD8217 features a factory-set gain of 20 V/V, with a maximum ±0.35% gain error over the entire temperature range. The initial offset, specified ±300 µV over temperature, and the miniscule temperature drift, ±100 nV/°C, will enhance any error budget. The buffered output voltage directly interfaces with any typical analog-to-digital converter. Regardless of the common mode, the AD8217 provides the correct output voltage when the input differential is at least 1 mV. Using a 10-mΩ shunt resistor, as above, the minimum current can be as low as 100 mA. The single-chip solution avoids the temperature drift and power consumption issues of the discrete solution.
Figure
4 is a linearity plot of the output voltage across R R
is 13 Ω;
_{G}R is 100 Ω;
_{BIAS}R_{1} is 10 kΩ;
load resistance is 200 Ω;
R is 200 Ω;
the Zener diode output is 5.1 V; the op amp is AD8628; the MOSFET
is BSS84. The maximum relative error is 0.69%, and the average
is 0.21% after calibration._{L}
Figure
5 is a linearity plot of the output voltage of the AD8210 as
a function of low-end values of input current flowing through
R R_{1}
is 20 kΩ;
R_{2} is 0.5 kΩ;
the load resistance is 200 Ω.
The external reference buffer is an AD8603. The maximum relative
error is 0.03%, and the average is 0.01% after calibration.
Figure
6 is a linearity plot of the output voltage of the AD8217 vs.
low-end values of input current flowing through R
Note that the tests were necessarily concentrated at the low end of the range and did not cover the full 50 mA to 20 A range. The reason is that the linearity challenge is mainly at the low output voltage (low unipolar current) portion of the range. Temperature experiments were also done for each solution at –40°C, +25°C, and +85°C. Table 2 shows the maximum relative error and average error when using the same correction factor at +25°C to calibrate the data at –40°C and +85°C.
If a temperature sensor is available for use in the system, different correction factors can be used to calibrate the data at different temperatures, but with increased component and manufacturing cost. Table 3 shows the maximum relative error and average error when different correction factors are used at –40°C, +25°C, and +85°C.
The temperature experiments show the high accuracy available over wide temperature ranges with devices using auto-zero technology, especially in the case of the AD8217.
Based on these test results, the AD8217 solution is the most suitable of the three choices for wide dynamic range unidirectional high-side current sensing and monitoring. We also note that an AD8210 solution offers operation right down to 0-V inputs—which could be beneficial for short-to-ground conditions. Note also that the AD8210 is capable of single-chip monitoring of bipolar current flow, as in charge/discharge applications. Calibration and temperature sensing are recommended in practical system designs requiring best system performance.
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