Design & Integration Files
- Bill of Materials
- Gerber Files
- Assembly Drawing
Part Numbers with "Z" indicate RoHS Compliance. Boards checked are needed to evaluate this circuit
- AD9434-500EBZ ($250.00) AD9434 evaluation board used to evaluate this circuit. Please see "Circuit Evaluation & Test" section for connection information.
Software such as C code and/or FPGA code, used to communicate with component's digital interface.
Features & Benefits
- 12-Bit, 500MSPS Wideband Receiver
- Antialiasing Filter
- Low Distortion Differential Driver
- 64dB SNR @ 140MHz Input
- 70dB SFDR @ 140MHz Input
Circuit Function & Benefits
The third-order Butterworth antialiasing filter is optimized based on the performance and interface requirements of the amplifier and ADC. The total insertion loss due to the filter network, transformer, and other resistive components is only 1.2 dB.
The overall circuit has a bandwidth of 290 MHz with a pass-band flatness of 1 dB. The SNR and SFDR measured with a 140 MHz analog input are 64.1 dBFS and 70.4 dBc, respectively.
The ADA4960-1 is an ideal driver for the AD9434, and the fully differential architecture through the low-pass filter and into the ADC provides good high frequency common-mode rejection, as well as minimizes second-order distortion products. The ADA4960-1 provides a gain of 0 dB to 18 dB, depending on the external gain resistor. In the circuit, a gain of 3.4 dB was used to compensate for the insertion loss of the filter network (1.1 dB) and the transformer (0.1 dB), providing an overall signal gain of 2.3 dB. An input signal of approximately 5.4 dBm produces a full-scale 1.5 V p-p differential signal at the ADC input.
The antialiasing filter is a third-order Butterworth filter designed with a standard filter design program. A Butterworth filter was chosen because of its flat response within the pass band. A third order filter yields an ac noise bandwidth ratio of 1.05 and can be designed with the aid of several free filter programs such as Nuhertz Technologies Filter Free(hwww.nuhertz/filter) or the Quite Universal Circuit Simulator (Qucs) Free Simulation (www.qucs.sourceforge.net).
In order to achieve best performance, the ADA4960-1 should be loaded with a net differential load of 100 Ω. The 5 Ω series resistors isolate the filter capacitance from the amplifier output, and the 62 Ω resistors in parallel with the downstream impedance yield a net load impedance of 101 Ω when added to the 10 Ω series resistance.
The 5 Ω resistors in series with the ADC inputs isolate internal switching transients from the filter and the amplifier. The 511 Ω resistor in parallel with the ADC serves to reduce the input impedance of the ADC for more predictable performance.
The third-order Butterworth filter was designed with a source impedance of 70 Ω, a load impedance of 338 Ω, and a 3 dB bandwidth of 360 MHz. The calculated values from the program are shown in Figure 2.
Figure 2. Design for 3rd Order Differential Butterworth Filter with ZS = 70 Ω, ZL = 338 Ω, FC = 360 MHz
The values chosen for the filter’s passive components were the closest standard values to those generated by the program.
The internal 1.3 pF capacitance of the ADC was subtracted from the value of the second shunt capacitor (10.01 pF) to yield a value of 8.71 pF. In the circuit, this capacitor was realized using two 18 pF capacitors connected to ground as shown in Figure 1. This provides the same filtering effect, as well as offering some ac common-mode rejection.
The measured performance of the system is summarized in Table 1, where the 3 dB bandwidth is 290 MHz. The total insertion loss of the network is approximately 1.1 dB. The bandwidth response is shown in Figure 3; the SNR and SFDR performance are shown in Figure 4.
Figure 3. Pass-Band Flatness Performance vs. Frequency
Filter and Interface Design Procedure
To achieve optimum performance (bandwidth, SNR, SFDR, etc.), there are certain design constraints placed on the general circuit by the amplifier and the ADC:
- The amplifier should see the correct dc load recommended by the data sheet for optimum performance.
- The correct amount of series resistance must be used between the amplifier and the load presented by the filter. This is to prevent undesired peaking in the pass band.
- The input to the ADC should be reduced by an external parallel resistor, and the correct series resistance should be used to isolate the ADC from the filter. This series resistor also reduces peaking.
The generalized circuit shown in Figure 5 applies to most high speed differential amplifier/ADC interfaces and will be used as a basis for the discussion. This design approach will tend to minimize the insertion loss of the filter by taking advantage of the relatively high input impedance of most high speed ADCs and the relatively low impedance of the driving source (amplifier).
Figure 5. Generalized Differential Amplifier/ADC Interface with Low-Pass Filter
The basic design process is as follows:
- Select the external ADC termination resistor RTADC so that the parallel combination of RTADC and RADC is between 200 Ω and 400 Ω.
- Select RKB based on experience and/or the ADC data sheet recommendations, typically between 5 Ω and 36 Ω.
- Calculate the filter load impedance using:
ZAAFL = RTADC || (RADC + 2RKB)
- Select the amplifier external series resistor RA. Make RA less than 10 Ω if the amplifier differential output impedance is 100 Ω to 200 Ω. Make RA between 5 Ω and 36 Ω if the output impedance of the amplifier is 12 Ω or less.
- Select RTAMP so that the total load seen by the amplifier, ZAL, is optimum for the particular differential amplifier chosen using the equation:
ZAL = 2RA + (ZAAFL || 2RTAMP)
- Calculate the filter source resistance:
ZAAFS = 2RTAMP || (ZO + 2RA)
- Using a filter design program or tables, design the filter using the source and load impedances, ZAAFS and ZAAFL, type of filter, bandwidth, and order. Use a bandwidth that is about 40% higher than one-half the sampling rate to ensure flatness in the frequency span between dc and fs/2.
- The internal ADC capacitance, CADC, should be subtracted from the final shunt capacitor value generated by the program. The program will give the value CSHUNT2 for the differential shunt capacitor. The final common-mode shunt capacitance is
CAAF2 = 2(CSHUNT2 − CADC)
After running these preliminary calculations, the circuit should be given a quick review for the following items.
- The value of CAAF2 should be at least 10 pF so that it is several times larger than CADC. This minimizes the sensitivity of the filter to variations in CADC.
- The ratio of ZAAFL to ZAAFS should not be more than about 7 so that the filter is within the limits of most filter tables and design programs.
- The value of CAAF1 should be at least 5 pF to minimize sensitivity to parasitic capacitance and component variations.
- The inductor, LAAF, should be a reasonable value of at least several nH.
In some cases, the filter design program may provide more than
one unique solution, especially with higher order filters. The
solution that uses the most reasonable set of component values
should always be chosen. Also choose a configuration that ends
in a shunt capacitor so that it can be combined with the ADC
Circuit Optimization Techniques and Trade-Offs
The parameters in this interface circuit are very interactive; therefore, it is almost impossible to optimize the circuit for all key specifications (bandwidth, bandwidth flatness, SNR, SFDR, gain, etc.). However, the peaking which often occurs in the bandwidth response can be minimized by varying RA and RKB.
The pass-band peaking is reduced as the value of the output
series resistance, RA, is increased. However, as the value of this resistance increases, there is more signal attenuation, and the amplifier must drive a larger signal to fill the ADC’s full-scale input range.
The value of RA also affects SNR performance. Larger values, while reducing the bandwidth peaking, tend to slightly increase the SNR because of the higher signal level required to drive the ADC full scale.
Select the RKB series resistor on the ADC inputs should be selected to minimize distortion caused by any residual charge injection from the internal sampling capacitor within the ADC. Increasing this resistor also tends to reduce bandwidth peaking.
However, increasing RKB increases signal attenuation, and the
amplifier must drive a larger signal to fill the ADC input range.
Another method for optimizing the pass-band flatness is to
vary the filter shunt capacitor, CAAF2, by a small amount.
The ADC input termination resistor, RTADC, should normally be
selected to make the net ADC input impedance between 200 Ω
and 400 Ω. Making it lower reduces the effect of the ADC input
capacitance and may stabilize the filter design but increases the
insertion loss of the circuit. Increasing the value will also reduce
Balancing these trade-offs can be somewhat difficult. In this
design, each parameter was given equal weight; therefore, the
values chosen are representative of the interface performance
for all the design characteristics. In some designs, different
values might be chosen to optimize SFDR, SNR, or input drive
level, depending on system requirements.
Note that the signal in this design is ac coupled with the 0.1 μF
capacitors to block the common-mode voltages between the
amplifier, its termination resistors, and the ADC inputs. Please
refer to the AD9434 data sheet for further details regarding common-mode voltages.
Passive Component and PC Board Parasitic Considerations
The performance of this or any high speed circuit is highly dependent on proper PCB layout. This includes, but is not limited to, power supply bypassing, controlled impedance lines (where required), component placement, signal routing, and power and ground planes. See Tutorial MT-031 and Tutorial MT-101 for more detailed information regarding PCB layout for high speed ADCs and amplifiers.
Low parasitic surface-mount capacitors, inductors, and resistors
should be used for the passive components in the filter. The
inductors chosen are from the Coilcraft 0603CS series. The
surface-mount capacitors used in the filter are 5%, C0G, 0402-
type for stability and accuracy.
See the CN-0238 Design Support Package (www.analog.com/CN0238-DesignSupport) for complete documentation on the system.
For applications that require less resolution, the 8-bit, 500 MSPS
AD9484 is pin compatible with the AD9434. The AD9484 has an SNR of 47 dBFS at 250 MHz analog input frequencies.
For applications that require a lower sampling rate, the 12-bit,
170 MSPS/ 210 MSPS/ 250 MSPS AD9230 is a pin-compatible ADC with approximately the same dynamic performance as the AD9434.
Also the 12-bit, 500 MSPS AD6641 could be considered for those applications that require digital predistortion (DPD) observation. This product has an on-chip 16k × 12-bit FIFO.